关键词: No-opto flyback, synchronous flyback converter, peak current mode controller
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应用笔记 6394

摘要 : This application note explains the procedure for designing a no-opto flyback converter with secondary-side synchronous rectification using the MAX17690 and MAX17606 to achieve high efficiency and better thermal management.

Using a flyback converter for low- and medium-power applications is the preferred design choice due to the flyback converter’s simplicity and low cost. However, in isolated applications, the use of optocoupler or auxiliary winding for voltage feedback across the isolation boundary increases the number of components, and design complexity. The MAX17690, a no-opto flyback controller, eliminates the optocoupler or auxiliary winding, and achieves ±5% output voltage regulation over line, load, and temperature variations.

In low output voltage and high output current applications, the diode on the secondary side of the flyback converter dissipates a significant amount of power; this power loss reduces the converter’s efficiency. The MAX17606, a secondary-side synchronous MOSFET driver, helps in replacing the secondary diode with a MOSFET. This improves the efficiency and simplifies thermal management.

This application note provides the step-by-step procedure for designing the different components of the MAX17690 + MAX17606-based synchronous flyback design.

The following specification is selected to demonstrate the design calculations for the MAX17690 and the MAX17606-based flyback converter. Figure 1 shows the typical application circuit for this application.

*Figure 1. Application circuit.*

Input voltage range | 18V to 36V |

Output voltage | 5V |

Maximum load current | 1A |

Steady-state output voltage ripple | 1% of output voltage |

Use the V_{INmin} and V_{INmax} from the selected specifications in the below equations to calculate the maximum duty cycle, D_{max}.

where:

V_{IN min} is the minimum input voltage in volts.

V_{IN max} is the maximum input voltage in volts.

D_{max} is the maximum operating duty cycle. If the calculated duty cycle is > 0.65, then choose D_{max} to be 0.65p.u.

For the present application, the switching frequency is selected as 150kHz. The R_{RT} is calculated for the selected f_{SW}.

Ω

Ω

A standard resistor of 33.2kΩ is selected.

The MAX17690 and the MAX17606 are specifically designed for the isolated flyback converters operating in Discontinuous Conduction Mode (DCM) or Border Conduction Mode (BCM). Use the below equations to select the transformer magnetizing inductance (L_{MAG}) for DCM operation.

For the present design L_{MAG} is selected to be 46.4uH, and the allowable tolerance on the L_{MAG} is ±10%. For the selected f_{sw} and the L_{MAG}, recalculate the D_{max} using the equations below:

=0.5p.u

The MAX17606 programs the turn-off trip point and decides the instant at which the secondary MOSFET is turned off. Due to the variation on the turn-off point, the secondary MOSFET conduction time changes. To guarantee the DCM operation of the converter for the variations on the turn-off threshold, magnetizing inductance (±10%), and the switching frequency (±6%), select the turns-ratio (K) based on the below equations:

=0.177

For the present design, K is chosen as 0.18 ±1%.For the selected L_{MAG} and f_{SW}, the primary peak current is calculated using the equation below:

=1.28A

The threshold voltage of the peak current-limit comparator is set at 100mV (typ) and 90mV (min). With the expected tolerance of ±10% on the L_{MAG} and the ±6% on the f_{SW}, to deliver the full-load power in all operating conditions, use the equations given below to calculate the current-sense resistor (R_{CS}) value.

= 62.5mΩ

A standard resistor of 62.5mΩ ±1% is selected.

The values of the resistor-divider can be selected so that the EN/UVLO pin voltage exceeds the 1.215V (typ) turn-on threshold at the desired input bus voltage (V_{START}). The same resistor-divider can be modified with an additional resistor (R_{OVI}) to implement input overvoltage(V_{OVI}) protection in addition to the EN/UVLO functionality, as shown in **Figure 1**. When the voltage at the OVI pin exceeds 1.215V (typ), the device stops switching. With the preselected value of 10kΩ for R_{OVI}:

For the present application V_{START} and V_{OVI} are selected to be 17.5V and 36.2V.

A standard resistor of 280kΩ is selected.

Since in this design the secondary MOSFET is always programmed to conduct at the sampling instant of the output voltage, there is no need to compensate the diode forward-voltage temp coefficient on the primary. For details on how to select the R_{TC} resistor for the other applications, refer to the MAX17690 IC data sheet.

The R_{IN}, R_{FB}, and the R_{SET} resistors program the output voltage and sampling instant for proper sampling of the output voltage. Use the below equations to calculate these values:

,and

Use the combination of standard resistors 274kΩ and 3.74kΩ to meet the required R_{FB} value of 277.7kΩ.

= 166.6kΩ

A standard resistor of 165kΩ is selected for this application.

In practice, due to the drop across the secondary leakage inductance of the transformer, the measured output voltage can deviate from the target output voltage. Use the below equations to readjust the output voltage to the desired value:

For the desired soft-start time(t_{SS} = 10ms), the SS capacitor is selected using:

= 50nF

The soft-start capacitor selected is 47nF for the present design.

The resistor connected between the VCM pin and SGND is used to scale the common-mode voltage of internal circuits within the operating range. Follow the below steps to select the R_{VCM} resistor value for proper operation.

- Calculate the internal scaling factor:
- From the below table, choose the row that has the equal or higher value for Kc with respect to the calculated Kc in Step 1. Select the row with Kc = 160 for the present design.
- Select the resistor value from the corresponding row as the R
_{VCM}(R_{VCM}=124kΩ).

= 111.1

Kc | RVCM (Ω) |
---|---|

640 | 0 |

320 | 75k |

160 | 124k |

80 | 220k |

40 | Open |

The “RCD and RC Snubber Circuit” section covers the selection of snubber components to limit the drain-to-source voltage to V_{DSmax} value selected in the above equation.

The RMS current in the MOSFET can be calculated using the below equation:

In the present application, the FDMS86252 part is selected as the primary MOSFET to achieve high efficiency. From the MOSFET data sheet R_{DS(ON)} value , the conduction loss in the MOSFET can be calculated using the equation given below:

From the MOSFET data sheet, the C_{OSS} at 100V is given as 60pF.

=50mW

It is important to verify the maximum junction temperature of the MOSFET for the calculated losses using the below equation. where TA is the ambient temperature, the R_{TH(JA)} is the MOSFET thermal resistance from junction-to-ambient, and the ^{P}MOSFET is the total MOSFET losses.

In this high-efficiency design, for the selected MOSFET the total losses are a very small portion of output power, and its junction temperature is within the limits.

Use the below equation to calculate the IC driver losses for the selected MOSFET:

For the secondary MOSFET, the RMS current equation is given below:

For the stable operation of MAX17690 + MAX17606-based designs over the entire operating conditions, it is recommended to select the R_{DS(ON)} of the secondary MOSFET such that the voltage across the MOSFET (at room temperature) is greater than the 100mV when the peak secondary current is flowing through the MOSFET.

In the present application, STL51N3LLH5 is selected as the secondary MOSFET.

The losses in the secondary MOSFET can be calculated using the loss equations provided in the primary MOSFET selection section and the maxim junction temperature can be verified to be within limits.

Use the following equations to calculate the snubber components:

where

The voltage rating of the snubber diode is:The RC component values are selected to be 60.4kΩ, 2.2nF.

*Figure 2. Waveforms with RCD clamp.*

*Figure 3. RC and RCD clamp circuitry.*

The RCD clamp only limits the maximum voltage stress on the primary MOSFET, but the ringing due to interaction between Llk and Cpar on the drain node is not damped. Because the MAX17690 uses the drain voltage information to sample the output voltage, it is important to damp this ringing within 350ns from the NDRV falling. In designs where this ringing is dominant, an RC snubber placed across the transformer primary winding damps this ringing. Use the following steps for designing an effective RC snubber:

- Measure the ringing time period from the drain node voltage.
- Add a test capacitance starting with 100pF until the time period of the ringing is 1.5 to 2 x t
_{1}. For the added capacitance C_{D}, measure the new ringing time period: - Use the following equation to calculate the drain node capacitance:
- Use the following equation to calculate the leakage inductance:
- Now, use the following equations to calculate the RC snubber values:

1.5 to 2 x the

The R_{c} and C_{c} values are selected to be 47Ω and 220pF.

Based on the actual ringing time (t

, where R_{TOFF} is in kΩ and t_{R} is in ns.

where:

L_{stray} is the MOSFET package lead inductance (see **Table 2** for lead inductance of various packages).

V_{trip}: V_{trip} should be selected as 0mV (corresponds to zero secondary current instant) for f_{sw} ≤ 100kHz and -6mV for f_{sw} > 100kHz. This ensures the proper output voltage sampling and stable operation of the MAX17690 + MAX17606-based design.

*Figure 5. Equivalent circuit of the MAX17606.*

S.No | Package | Stray inductance (nH) |
---|---|---|

1 | DFET | 0.5 |

2 | MLP | 1.8 |

3 | SO-8, PowerPAK^{®} |
1.8 |

4 | DPAK | 2.7 |

5 | D2PAK | 5.2 |

In practice, due to delay from the comparator circuit and MOSFET turn-off time (t_{OFF}, given in the MOSFET data sheet), the R_{DRN} equation given above does not predict the exact turn-off instant. The equation given below includes these delays and determines the turn-off instant:

where:

, t_{DELAY} can be calculated using **Table 3**.

S.No | (mV/μs) |
t_{DELAY}(ns) |
---|---|---|

1 | 100.00 | 41 |

2 | 66.67 | 45 |

3 | 44.44 | 47 |

4 | 29.63 | 53 |

5 | 19.75 | 56 |

6 | 13.17 | 63 |

7 | 8.78 | 65 |

8 | 5.85 | 80 |

For the present design:

mV

= 2.47kΩ

A standard resistor of 2.49kΩ is selected to be the DRN resistor.

The MAX17690 offers a hiccup scheme that protects and reduces power dissipation in the circuit under output short-circuit conditions. One occurrence of the runaway current limit, or output voltage less than 70% of regulated voltage, would trigger a hiccup mode that protects the converter by immediately suspending the switching for the period of 16,384 clock cycles. The threshold voltage of the runaway current limit comparator is set at 120mV (typ).

The MAX17690 samples the output voltage feedback when the primary MOSFET is turned off and energy stored during the “ON time” is being delivered to the secondary. Therefore, it is mandatory to switch the external MOSFET to sample the reflected output voltage. Due to the default switching, a minimum amount energy is delivered to the output capacitor under no-load conditions. This small minimum load can easily be provided on the output by connecting a fixed resistor. In the absence of a minimum load, or a load less than the “minimum load,” the output voltage rises to higher values. To protect for this condition, a Zener diode of appropriate breakdown voltage rating can be installed on the output. Care should be taken to ensure that the Zener breakdown voltage is outside the output voltage envelope in both steady-state and transient conditions.

Under ideal circuit working conditions, the MAX17690 is designed to regulate the output voltage with 1% of full-load rated current on the output. With nonidealities, in most of the designs the current required to regulate the output voltage is less than 2% of the full-load rated current.

**Note**: Refer to the MAX17690 IC data sheet for more information.

A Zener diode with a Zener breakdown of 10% to 15% higher than the output voltage can serve as a minimum load if preloading is not acceptable. For a 5V output voltage, the Zener breakdown (V_{ZenerBR}) is selected to be 5.6V. The maximum power dissipation in the Zener diode at no-load is calculated as:

where I_{minload} is the minimum load required.

In the present design, the 2% of full-load current is 20mA.

For the present design, a 5.6V, 0.5W MMSZ5232B Zener is selected. The resistor in series with the Zener is calculated based on the Zener breakdown voltage and the desired no-load output voltage.

For the present design, the output voltage at absolute no load is set at 6V.

A standard resistor of 22Ω is selected.

The power dissipation in this resistor is given by:

Considering a 2% ripple on the minimum supply voltage, the input capacitance is:

=3.3μF

Two 2.2μF, 100V 1210 capacitors have been used in the present design considering the DC-biasing.

From the DC-bias characteristics, the 100μF, 6.3V 1210 capacitor offers 43mF at 5V. Hence, two 100μF, 6.3V 1210 capacitors are selected for the present design.

The output voltage ripple is determined by the bulk capacitance and ESR (R_{ESR}) of the output capacitor. When using ceramic capacitors, the ESR ripple can be neglected in most of the cases. For the high-ripple current aluminum capacitor, the capacitance calculation begins with the maximum acceptable ripple voltage and how this ripple should be divided between the ESR step and the ripple offered by the bulk capacitance.

For a 1% contribution to the total ripple voltage, the ESR of the output capacitor should be:

For a 1% contribution to the total ripple voltage, the bulk capacitance should be:

Load pole ==740.1Hz

R_{z} = 4.39kΩ

A standard resistor of 4.3kΩ is selected.

A standard capacitor of 47nF is selected.

== 493pF

A standard capacitor of 470pF is selected.

**Note:** When the ESR zero of the output capacitor is significant, the compensator pole capacitor (C_{p}) should be selected to cancel the ESR zero.

- Keep the loop area of paths carrying the pulsed currents as small as possible. In flyback design, the loop created by the V
_{IN}bypass capacitor, transformer primary winding, MOSFET switch, and sense resistor is critical. Similarly, the high-frequency current path for the MOSFET gate switching from the INTVCC capacitor through the source of the MOSFET and sense resistor is also critical. - The INTVCC bypass capacitor should be connected right across the INTVCC and PGND pins of the MAX17690.
- A bypass capacitor should be connected across the V
_{IN}and SGND pins and should be placed close to MAX17690. - The IC’s exposed pad should be directly connected to the MAX17690’s SGND pin. The exposed pad should also be connected to the SGND plane in other layers by means of thermal vias under the exposed pad so that the heat flows to the large “signal ground” (SGND) plane.
- The R
_{FB}resistor trace length should be kept as small as possible. - The PGND connection from the INTVCC capacitor and the SGND plane should be star connected at the negative terminal of the current-sense resistor.
- The proper sensing of drain-to-source voltage across the secondary MOSFET is critical in MAX17606. The R
_{DRN}should be Kelvin connected to the drain of the synchronous MOSFET. The source pin of the MOSFET should also be Kelvin connected to the MAX17606 GND pin. - Connect the R
_{TOFF}resistor directly between TOFF pin and the MAX17606 GND pin. The return path should not be connected to ground plane.