
关键词: resonant reset, forward converter, singletransistor, sinusoidal reset, synchronousrectifier
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Designing SingleSwitch, ResonantReset, Forward Converters
By: 
Suresh Hariharan, Director and Product Definer 

Abstract: Among powerconverter topologies, the singletransistor, forward converter is one of the most common for power levels below 100 watts. This article describes an improvement to that circuit called the "singletransistor, resonantreset, forward converter," which eliminates the reset winding and a diode (D_{TR}). Several other advantages of this design will be discussed.
A similar article appeared in the October 2005 issue of
Power Electronics Technology.
Introduction
Singletransistor, resonantreset forward converters are commonly used in DCDC converter modules for power levels below 100 watts. These devices are also quite useful for DCDC converters with widely adjustable output voltages. This article, however, describes an improved circuit called the "singletransistor resonantreset forward converter." This design eliminates the reset winding and a diode (D
_{TR}), and offers several distinct advantages.
The duty cycle for this resonantreset converter can exceed 50%, making it suitable for lowcost DCDC converters that operate from wide input voltages and deliver widely varying outputs. The absence of a reset winding reduces costs by simplifying the transformer, especially for the planar transformers widely used in highdensity DCDC converter modules. Finally, the resonantreset circuit's sinusoidal reset voltage reduces EMI.
Conventional SingleSwitch Forward Converter Design
To properly appreciate the resonantreset topology, we must first understand the conventional singleswitch forward converter (
Figure 1a). When switch Q1 turns on, the transformer current rises from zero and the diode, D
_{TR}, is reverse biased. Transformer magnetizing current builds up to a value I
_{M} = V
_{IN}T
_{ON}/L
_{M}, where T
_{ON} is the ON time per switching cycle and L
_{M} is the magnetizing current.
Figure 1a. Conventional singletransistor forward converter.
During the switch ON period, the load current, I
_{O}, is reflected in the primary as I
_{P} = I
_{O}N
_{S}/N
_{P}, where N
_{S} is the number of secondary turns and N
_{P} is the number of primary turns. Output voltage is V
_{O} = V
_{IN}DN
_{S}/N
_{P}, where D = T
_{ON}/T
_{S} and 1/T
_{S} is the switching frequency. Magnetizing current in the transformer primary just before turnoff is V
_{IN}T
_{ON}/L
_{M}. When Q1 turns off, the transformer voltage tends to reverse. Voltage on the D
_{TR} cathode keeps increasing until D
_{TR} turns on.
For typical applications the N
_{P}/N
_{R} turns ratio is 1, where N
_{R} is the number of turns in the primary reset winding. The transformer magnetizing current now decreases from I
_{M} to zero. When it reaches zero, the transformer is fully reset and voltage across the transformer remains at zero until the next switching cycle starts. The maximum duty cycle, D
_{MAX}, in these applications is limited to 50%.
ResonantReset Forward Converter Design
Singleswitch resonantreset forward converters are characterized by the absence of a reset winding (
Figure 1b). During the OFF time, the transformer resets (without loss) through a resonant circuit that consists of: the magnetizing inductance; and the combined capacitance of the switch, primary winding, and all reflected secondary capacitances including the rectifyingdiode capacitance.
Figure 1b. Singleswitch, resonantreset forward converter.
Description of Operation
The following assumptions are made for this circuit analysis:
 The circuit has reached steadystate operation.
 L_{O} and C_{O} (fairly large) can be considered infinite.
 Leakage inductance is neglected.
 Drops due to the diode and switch onresistance are neglected.
Steadystate operation for the circuit comprises three intervals in each switching cycle:
Interval 1
Initially, t = 0 and Q1 is ON (
Figure 2a). The transformer is magnetized with a ramp current during the switch ON period. Secondary current flows through the secondary diode, D
_{R}, and the voltage across capacitance, C
_{D}, is approximately zero. C
_{D} includes the internal diode capacitance and the external capacitance added across D
_{R}. The primary magnetizing current has a value I
_{1} at the start of this interval, and I
_{2} at the end of the interval.
Primary current is the sum of the reflected current, I
_{O}(N
_{S}/N
_{P}), and the primary magnetizing current.
Figure 2a. From Figure 1b, an equivalent circuit for the voltage on Q1 and the primary magnetizingcurrent waveform during Interval 1 (not to scale).
Interval 2
When the switch is turned off, the switch draintosource voltage begins to rise (
Figure 2b). When that voltage exceeds V
_{IN}, the secondary diode, D
_{R}, turns off and the freewheeling diode, D
_{F}, turns on. A sinusoidal demagnetization current starts to flow through the resonant circuit that is formed by the parallel combination of transformer magnetizing inductance, L
_{M}, and the capacitance, C
_{R}, reflected across the transformer primary. That capacitance, C
_{R}, is the sum of all capacitances across the primary, including the reflected diode capacitance, C
_{D}:
where C
_{S} is the primary switch capacitance and C
_{T} is the transformer primary capacitance. C
_{D} is the external capacitance across diode C
_{D} (diode capacitances << C
_{D}). Interval 2 ends at the end of T
_{ON} + T
_{R}, where T
_{R} is one half of a resonant interval.
The external capacitance, C
_{R}, charges from zero to a peak value of
during this interval, and then discharges back to zero. The magnetizing current, I
_{1}, at the end of the interval should therefore equal I
_{2}. Voltage on the primary switch at the end of this interval is V
_{IN}, but voltage on the switch reaches a peak of
halfway through the interval.
Figure 2b. From Figure 1b, an equivalent circuit for the voltage on Q1 and the primary magnetizingcurrent waveform during Interval 2 (not to scale).
Interval 3
During this interval the diodes D
_{R} and D
_{F} are both ON; the primary switch is OFF (
Figure 2c). Voltage across the transformer is zero, and the magnetizing current at the end of the interval equals I
_{2}. This ends a switching cycle. Because the circuit is at steady state, the current I
_{1} therefore equals I
_{2}. Substituting for I
_{1} in equation 1, we see that the primary magnetizing current at the start of each switching cycle is:
The primaryswitch voltage remains at V
_{IN} during Interval 3. Note at the end of T
_{S}, that I
_{2} ≠ I
_{1} is possible if
. In that case a full halfcycle of resonance has not been completed before the next switching cycle begins. Therefore the voltage across the primary switch exceeds V
_{IN} at the start of each switching cycle. That condition increases the switching loss.
Figure 2c. From Figure 1b, an equivalent circuit for the voltage on Q1 and the primary magnetizingcurrent waveform during Interval 3 (not to scale).
Transient Operation
Transient stresses on the primary switch and secondary output diodes can vary greatly, depending on the type of controller used in the application. If the design is not optimal, transients can cause failure in the primary switches or the secondary diodes.
Consider operation with a currentmode PWM controller. Initially, the power supply operates at no load and high line voltage. A load transient is applied (minimum load to full load), which causes an immediate dutycycle step to maximum duty cycle. In turn, that event causes a large increase in the transformer's magnetizing current, and may saturate the transformer unless its design accounted for such transients. The resonantreset voltage is much higher than that during steadystate operation, and can cause failure in the forward diode or the primary switch.
To combat this problem, we introduce a voltmicrosec clamp. Consider the controller above with a maximumdutycycle clamp that is inversely proportional to the input voltage. That arrangement limits the maximum flux excursion along the transformer's BH loop during a transient, which in turn allows the use of a smaller transformer. Transientvoltage stress on the forward diode and the primary switch is significantly less, but is still higher than during steadystate operation.
Now consider the operation of this converter type with a very light load, and using diodes for rectification. Magnetizing current is very close to zero during this mode of operation, and the duty cycle is low. If we now apply a load transient (from no load to full load), the duty cycle immediately increases to the maximum value allowed by the adaptive dutycycle clamp. Before application of the transient, the magnetizing current is zero. The transient peak duty cycle at high line voltage is
, where V
_{INMIN} is the lowline input voltage, D
_{MAX(TR)} is the maximum duty cycle at low line voltage set by the adaptive dutycycle clamp, and V
_{INMIN} is the input voltage at high line voltage. When a transient occurs, the magnetizing current increases from 0 to
in the first switch ON cycle after the transient. Here L
_{M} is the primary magnetizing inductance and ƒ
_{SW} is the switching frequency. After the switch turns off, the magnetizing current reverses in a sinusoidal fashion set by the magnetizing inductance, L
_{M}, and capacitance, C
_{R}. Peak voltage on the switch is:
For steadystate operation at full load and high line voltage, the peak steadystate voltage on the switch is:
where D
_{MAX(S)} is the steadystate duty cycle at full load and low line. In practical applications we try to set D
_{MAX(TR)} slightly higher than D
_{MAX(S)}. We also see that the peak transient reverse voltage on the diode D
_{F} is more than twice as high as the peak steadystate reverse voltage with this type of PWM controller. For PWM controllers without the voltmicrosec clamp, the transient voltage can be even higher.
If the circuit includes synchronous rectifiers, the inductor current does not become discontinuous and the magnetizing currents at light load and at full load are almost the same. For PWM currentmode controllers with voltmicrosec clamps, the transientvoltage stress on the primary switch and the secondary diode, D
_{F}, is closer to the peak steadystate voltage stress.
The behavior of voltagemode controllers is similar to that of currentmode PWM controllers. Again, the use of an adaptive voltmicrosec clamp can reduce stress. These converter types often include a dutycycle softstart that ramps up the duty cycle, thus controlling any buildup of magnetizing energy while alleviating voltage stress.
Design Example
The working power supply of
Figure 3 accepts DC input voltages in the range 36V to 56V, and produces a 4V to 18V isolated variable output voltage, controlled by an adjustable external reference. The maximum output current is 0.4A and the switching frequency, ƒ
_{SW}, is 500kHz.
More detailed image (PDF, 212.38kB).
Figure 3. Resonantreset forward converter with an input range of ground to 48V_{OUT} (36V to 56V) and output range 4V to 18V.
The resonantreset forward converter is most suitable for this design because it lets us maximize the duty cycle. That capability is necessary if the output voltage is to be properly controlled from high levels down to 4V. Otherwise, the PWM controller's minimum ON time is a limitation that can introduce problems. Synchronous rectifiers should be included to maximize efficiency and enable the PWM controller to control the output voltage down to 4V at light loads. The currentmode PWM controller shown also includes an adaptive voltmicrosec clamp.
Adaptive DutyCycle Clamp
Because the power supply must turn on and provide full power at 36V, we set its turnon point at 34.2V. That turnon voltage includes a 5% margin to compensate for component tolerances. We then set the maximum duty cycle that corresponds to the turnon point (set by the adaptive duty cycle) at 75%. That approach leaves 25% of the switching time available for resetting the transformer at the converter's lowest operating voltage.
Primary MOSFET Voltage Rating
At the lowest operating voltage, the maximum available reset time for the transformer is:
where D
_{MAX} = 0.75 and ƒ
_{SW} = 5 x 10
^{5}. These values yield a reset time of 0.5µs. To minimize switching loss, the magnetizing current should complete one halfcycle of sinusoidal "resonant ringing" as given by Equation 4. Therefore,
, and the peak steadystate voltage stress on the primary switch (obtained by substituting values in equation 7) is 208.6V. Thus, for this design we choose a switch rated at 250V.
Transformer Design
Primarytosecondary turns ratio for the transformer is n:
We choose a transformer with an EFD15 core of 3F3 material, and obtain n
< 1.35 by substituting values in Equation 9. The actual primary turns (30) and secondary turns (24) yield a turns ratio of 1.25. The magnetizing inductance for this transformer, wound using ungapped cores, is 702µH ±25%. Tolerance in the magnetizing inductance could produce a tolerance of +11%/13.4% in the transformer's selfresonant frequency, not accounting for tolerance in the total capacitance appearing across the primary in the actual circuit. The measured selfresonant frequency of a sample transformer was lower than 1MHz.
We must guarantee that the actual circuit's demagnetizing selfresonant frequency is higher than ƒ
_{SW}/(1  D
_{MAX}). We therefore gap the core, both to reduce the transformer's measured selfresonant frequency and to reduce the variation in magnetizing inductance. Using a gapped core with A
_{1} tolerance of 10% yields an inductance of 144µH.
The selfresonant frequency measured for the new transformer sample is 4MHz; the transformer capacitance calculated from the expression for selfresonant frequency is 11pF. Based on the available reset time, the maximum allowable primary capacitance is 176pF. That latter value allows a maximum of 165pF for the sum of switch capacitance and reflected diode capacitance, C
_{R}. Because MOSFET capacitance is not easily determined, we must build the circuit and adjust the value of added capacitance across the synchronous MOSFET, Q
_{R}, to get the appropriate reset time. In the actual power supply, the added capacitance across MOSFET Q
_{R} is 100pF.
Output Inductor and Capacitor
The output inductor and capacitor are chosen to optimize efficiency and ensure compliance with the outputripple specification. Thus, the inductor value is 47µH, and C
_{O} is formed by connecting three ceramic capacitors in parallel, each rated 4.7µF and 25V.
Primary MOSFET
For the primary MOSFET, Q
_{1} (voltage rating of 250V), we choose an FQD4N25 from Fairchild for its low inherent capacitance and low onresistance. This MOSFET also minimizes the gatedrive loss, conduction loss, and switching loss.
SynchronousRectifier MOSFETs
Peak stress on the synchronous rectifier, Q
_{R}, is:
where n
_{a} is the power transformer's actual primarytosecondary turns ratio. In this case, n
_{a} is 1.25 and the calculated value of V
_{QR} is 122V. We therefore choose a 150V MOSFET for Q
_{R}. The peak voltage stress on the freewheeling MOSFET, Q
_{F}, is:
where n
_{a} is 1.25 and V
_{INMAX} is 56V. The calculated value is 44.8V, so for Q
_{F} we choose a MOSFET rated at 60V. (The control circuit and synchronous MOSFET drives are shown in the schematic, but not discussed further.)
Experimental Results
Figures 4,
5, and
6 show voltage waveforms on the primary MOSFET of Figure 3 at different input voltages and various output voltages, and with an output load of 400mA. The drainvoltage waveforms clearly show that the resonantreset voltage does not vary with line voltage, but is proportional to the output voltage. Peak voltage on the primary MOSFET is equal to the input voltage plus the resonantreset voltage.
Figure 4. From Figure 3, V_{DS} on Q14 at an input of 48V_{DC}, with output voltage at 4V (a) and 8V (b).
Figure 5. From Figure 3, V_{DS} on Q14 at an input of 48V_{DC}, with output voltage at 12V (a) and 18V (b).
Figure 6. From Figure 3, output voltage at 18V, with V_{DS} on Q14 at an input of 36V_{DC} (a) and 56V_{DC} (b).
Conclusion
Resonantreset forward converters are quite suitable for power supplies operating from widerange DCvoltage inputs. They are also suitable for applications requiring a wide range of adjustable output voltage. In designing resonantreset forward converters, you should minimize the stress of transient voltages on the devices; using synchronous rectification reduces transientvoltage stress on the power semiconductors. For optimum performance you should also choose an appropriate controller.
© Mar 20, 2007, Maxim Integrated Products, Inc.

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