
Keywords: rf, rf design, radio frequency, RF, Intercept point, IP, intermodulation, IM, continuous wave, CW, IIP2, IIP3, IM, IIM3, carriertointerference ratio, C/I
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Use Selectivity to Improve Receiver Intercept Point
© May 01, 2002, Maxim Integrated Products, Inc.


Abstract: A receiver's intermodulation (IM) spurious response attenuation is a measure of the receiver's ability to receive a modulatedinput RF signal frequency on its assigned channel frequency in the presence of two interfering continuous wave (CW) tones. The receiver's protection against spurious response interference is a measure of the its ability to discriminate between the input signal at the assigned frequency and an undesired signal at any other frequency to which it is responsive. This application note analyzes the sources of RF interference that affect receiver linearity. A dualconversion receiver is examined for offchannel interference response.
A similar version of this article appeared in the December 1997 issue of
RF Design magazine.
Introduction
A receiver's second and thirdorder intercept points (IPs) represent the two most important linearity specifications for a radio circuit or system. The intercept points allow us to predict the receiver's intermodulation (IM) performance, which describes the radio's susceptibility to interference from adjacent or nearby users. This article presents a modification of the traditional receiver intercept point cascading equation for both the second and thirdorder IM cases. The mathematical derivation for each of the secondorder (IP2) and thirdorder (IP3) intercept point cascading equations incorporates the effect of adding selectivity (S) between receiver stages to improve IIP2 and IIP3.
Note: in this article all variables in uppercase letters refer to dB or dBm units and all variables in lowercase letters refer to linear units.
Spurious Response Interference
In wireless specifications related to the recommended minimum performance standards for the mobile station, the receiver's IM performance is specified technically under two main topics: the receiver's IM spurious response attenuation, and the receiver's protection against spurious response interference.
The IM spurious response attenuation is a measure of the receiver's ability to receive a modulatedinput RF signal frequency on its assigned channel frequency in the presence of two interfering continuouswave (CW) tones. These tones are separated from the assigned input signal frequency so that the nthorder mixing of the two undesired signals can occur in the nonlinear elements of the receiver, thereby producing a third signal in the band of the desired signal. The receiver's protection against spurious response interference is a measure of its ability to discriminate between the input signal at the assigned frequency and an undesired signal at any other frequency to which it is responsive.
Interference from ThirdOrder IM Products
As a result of thirdorder mixing in a receiver's frontend, two offchannel CW tones at frequencies f
_{1} and f
_{2} introduce a thirdorder IM product at a frequency equal to (2f
_{1}  f
_{2}), which can appear in the onchannel signal passband (
Figure 1a). This inband thirdorder intermodulation (IM3) product reduces the carriertointerference ratio (C/I) at the receiver's demodulator. Based on a linear 3:1 slope (
Figure 1b), the level of this input IM3 product (IIM3, in dBm) can be determined using an equation that includes the receiver's overall input IP3 (IIP3, in dBm) and the input power level of each of the two offchannel CW tones (P
_{I}, in dBm):¹
IIM3 = 3 × P_{I}  2 × IIP3 (dBm) 
(Eq. 1) 
Figure 1. IM3 product as an inband interferer due to two offchannel CW tones (a), and thirdorder intercept point (IP) definition (b).
Figure 2 represents a conventional dualconversion superheterodyne receiver architecture. In this type of receiver architecture, the IM3 products from the offchannel CW interferers are generated in the lownoise amplifier (LNA), the first mixer, the IF amplifier, the second mixer, and the IF limiting amplifier. All the IM3 products merge at the demodulator input, resulting in an equivalent inband IM3 product at the receiver input (IIM3). This IM3 product, acting as an inband interferer, can be reduced by minimizing the IM3 contribution from the IF amplifier, the second mixer, and the IF limiting amplifier. The IM3 contribution is reduced by introducing a certain amount of IF selectivity (S) to these offchannel interferers in the IF filter following the first mixer (IF filter 1). Note that filter selectivity (S) refers to the attenuation in the IF filter 1 rejection band at the offchannel interferers' frequencies relative to the onchannel filter passband insertion loss (IL). Thus, the total rejection (R) (in dB) in the IF filter stopband at the offchannel CW tones' frequencies can be defined as: R = (IL + S). The IF filter selectivity (S) reduces the thirdorder distortion and dynamic range requirements of the subsequent receiver blocks. As a result, the overall receiver's IIP3 can be optimized in order to minimize the equivalent inband IIM3 product and thus meet the required receiver baseband (C/I) ratio.
Figure 2. Conventional superheterodyne dualconversion receiver.
Modified ThirdOrder Input Intercept Point (IIP3) Cascading Equation
In
Figure 3 the dualconversion receiver architecture (Figure 2) is divided into three blocks: the RF block, the IF filter 1, and the IF block. The RF block, or block 1, consists of the receiver RF stages preceding the first IF filter. The IF block, or block 2, consists of the receiver's IF stages following the first IF filter. Block 1 has a gain, G
_{1}, at RF and an equivalent thirdorder input intercept point IIP3
_{1}. Block 2 has a gain, G
_{2}, at IF and an equivalent thirdorder input intercept point IIP3
_{2}. The power level of each of the two offchannel CW interferers present at the receiver's input is assumed to equal P
_{I}, which is the level of the two offchannel CW tones' input to block 1. P
_{2} is the level of the two offchannel CW tones translated to IF and input to block 2. IIM3 is the total IM3 distortion power from the two offchannel CW tones that is referred to the receiver input. IIM3
_{1} is the total IM3 distortion power generated in block 1 that is referred to its input. IIM3
_{2} is the total IM3 distortion power generated in block 2 that is referred to its input.
Figure 3. Block diagram for deriving the modified IP3 cascading equation. The equation incorporates the effect of adding selectivity (S) to receiver stages at the two offchannel CW tones' frequencies. Power symbols are in dBm, and gain is in dB.
In the following derivation the input IM3 distortion voltage of block 1 and the input IM3 distortion voltage of block 2, divided by the voltage gain of preceding stages, are added exactly inphase as a worstcase condition. In this way we obtain the total IM3 distortion voltage that is referred to the receiver input. Assuming a system characteristic impedance of 1Ω, we can write the following:
√iim3 = √iim3_{1} + √(iim3_{2}/(g_{1}/il)) (volts) 
(Eq. 2) 
Where the square root converts the IM3 powers to voltages. The terms iim3, iim3
_{1}, and iim3
_{2} are in linear power units (watts or milliwatts). Also, G
_{1} (dB) = 10 × log
_{10}(g
_{1}) and IL (dB) = 10 × log
_{10}(il).
Equation 1 can be rearranged into the following equation:
IIP3 = P_{I} + ½(P_{I}  IIM3) (dBm) 
(Eq. 3) 
Equation 3 defines the input IP3 for the whole receiver, and can be rewritten in linear power units (milliwatts, or mW) instead of dBm:
p_{I}/iip3 = √(iim3/p_{I}) 
(Eq. 4) 
Similar to what we did in equation 3, we can define IIP3
_{1} and IIP3
_{2} for block 1 and block 2, respectively, as:
IIP3_{1} = P_{1} + ½(P_{1}  IIM3_{1}) (dBm) 
(Eq. 5) 
IIP3_{2} = P_{2} + ½(P_{2}  IIM3_{2}) (dBm) 
(Eq. 6) 
Knowing that P
_{1}(dBm) = P
_{I} and P
_{2}(dBm) = P
_{I} + (G
_{1}  IL  S), we obtain from equations 5 and 6:
IIP3_{1} = P_{I} + ½(P_{I}  IIM3_{1}) (dBm) 
(Eq. 7) 
IIP3_{2} = (P_{I} + G_{1}  IL  S) + ½(P_{I} + G_{1}  IL  S  IIM3_{2}) (dBm) 
(Eq. 8) 
As we have done for equation 3, equations 7 and 8 can be rewritten in linear power units instead of dBm to result in equations 9 and 10, respectively:
p_{I}/iip3_{1} = √(iim3_{1}/p_{I}) 
(Eq. 9) 
p_{I}(g_{1}/il)/(iip3_{2} × s^{3/2}) = √iim3_{2}/((g_{1}/il)p_{I}) 
(Eq. 10) 
Where S(dB) = 10 × log
_{10}(s) and IL(dB) = 10 × log
_{10}(il). Note that S(dB) and IL(dB) are positive numbers.
Returning to equation 2 and dividing it by (p
_{I})
^{1/2}, we get equation 11:
√(iim3/p_{I}) = √(iim3_{1}/p_{I}) + √(iim3_{2}/(g_{1}/il)p_{I} 
(Eq. 11) 
If we substitute each term in equation 11 by equivalent terms using equations 4, 9, and 10, and after simplifying by removing the term p
_{I}, we obtain the following modified IIP3 cascading equation:
1/iip3 = 1/iip3_{1} + (g_{1}/il)/(iip3_{2} × s^{3/2}) 
(Eq. 12) 
As we can see from equation 12, by choosing a highly selective IF filter (s >> 1), we can minimize the effect of the IF block input IP3 (IIP3
_{2}) on the overall receiver input IP3 (IIP3), which becomes almost defined by the RF block IIP3 (IIP3
_{1}). It is worth noting that in a cascaded system analysis the IF block input IP3 (IIP3
_{2}) should be replaced with an equivalent input IP3. This exchange effectively introduces selectivity ahead of the IF block. This equivalent IIP3
_{2} can be written as:
IIP3^{e}_{2} = IIP3_{2} + (3/2) × S (dBm) 
(Eq. 13) 
A more generalized equation that predicts the overall input IP3 of a receiver chain with M cascaded stages can be formulized based on equation 12. Each stage has a linear gain (g
_{n}), an input IP3 (IIP3
_{n}, in watts), and a selectivity (s
_{n}) at the two offchannel CW tones' frequencies. Together this introduces an inband IM3 product (assuming that il
_{n} << s
_{n}):
1/iip3 = 1/iip3_{1} + (g_{1}/(iip3_{2} × s_{1}^{3/2}) + (g_{1} × g_{2})/(iip3_{3} × (s_{1} × s_{2})^{3/2}) + ... + (g_{1} × g_{2} ... g_{M1})/(iip3_{M} × (s_{1} × s_{2} ... s_{M1})^{3/2}) 
(Eq. 14) 
Where S
_{n}(dB) = 10 × log
_{10}(s
_{n}). Note that this equation simplifies the classic equation for the intercept point calculation of a cascade of M stages when setting s
_{n} to 1. This translates into setting selectivity S
_{n} to 0dB.¹
Interference from SecondOrder IM Products
Receiver spurious responses are frequencies that differ from the onchannel RF signal frequency, yet they still produce an output interferer in the receiver passband if encountered at a sufficiently high level. One of these spurious response frequencies is at the halfIF frequency. This halfIF spurious response results in a secondorder IM product (IM2), which occurs in the receiver's RF frontend. The IM2 level can be predicted from the secondorder intercept point (IP2) of the receiver's RF frontend, which is defined up to and including the first mixer in the receiver chain (Figure 2).
For highside injection in the first mixer (
Figure 4a), a CW tone at the receiver's input, which is offset from the LO frequency by f
_{IF}/2, is downconverted to IF by the (2.f
_{CW} + 2.f
_{LO}) IM product that occurs in the first mixer.¹
^{,}² Similarly for lowside injection, a CW tone, which is offset from the LO frequency by +f
_{IF}/2, is downconverted to IF by the (2.f
_{CW}  2.f
_{LO}) IM product. Based on a linear 2:1 slope (
Figure 4b), the level of this input IM2 product (IIM2, in dBm) can be determined using an equation that includes the receiver's RF frontend input IP2 (IIP2, in dBm) and the power level of the input halfIF CW tone (P
_{I}, in dBm):¹
IIM2 = 2 × P_{I}  IIP2 (dBm) 
(Eq. 15) 
Figure 4. IM2 product as an inband interferer due to halfIF spurious response (a), and secondorder intercept point (IP) definition (b).
This inband IM2 product from the halfIF spurious response can be reduced by minimizing the secondorder IM contribution from the first mixer. This is done by introducing a certain amount of RF selectivity (S) to this offchannel interferer in the RF filters preceding the first mixer (RF filters 1 and 2). Note that filter selectivity (S) refers to the attenuation in the RF filter rejection band at the spurious response frequency relative to the onchannel filter passband insertion loss (IL). The RF filter selectivity (S) reduces the secondorder distortion and dynamicrange requirements of the first mixer. As a result, the overall receiver's RF frontend IIP2 can be optimized to minimize the equivalent inband IIM2 product from the halfIF tone, thus meeting the required receiver baseband (C/I) ratio.
Modified SecondOrder Input Intercept Point (IIP2) Cascading Equation
In
Figure 5, the dualconversion receiver RF frontend is divided into three blocks: RF filter 2, block 1 (which consists of all stages preceding RF filter 2), and block 2 (which follows RF filter 2 and includes the first mixer). Block 1 has a gain, G
_{1}, at RF and an equivalent secondorder input intercept point IIP2
_{1}. Block 2 has a gain, G
_{2}, at RF and an equivalent secondorder input intercept point IIP2
_{2}. The power level of each halfIF CW tone present at the receiver's input is assumed to be equal to P
_{I}, which is the level of the halfIF CW tone input to block 1. P
_{2} is the level of the halfIF CW tone input to block 2. IIM2 is the total IM2 distortion power from the halfIF CW tone that is referred to the receiver input. IIM2
_{1} is the total IM2 distortion power generated in block 1 that is referred to its input. IIM2
_{2} is the total IM2 distortion power generated in block 2 that is referred to its input.
Figure 5. Block diagram for deriving the modified IP2 cascading equation, which incorporates the effect of adding RF selectivity (S) to the receiver's RF frontend stages at the halfIF spurious frequency. Power symbols are in dBm, and gain is in dB.
In the following derivation the input IM2 distortion voltage of block 1 and the input IM2 distortion voltage of block 2, divided by the voltage gain of preceding stages, are added exactly inphase as a worstcase condition. In this way we obtain the total IM2 distortion voltage that is referred to the receiver's input. Assuming a system characteristic impedance of 1Ω, we can write the following:
√iim2 = √iim2_{1} + √(iim2_{2}/(g_{1}/il)) (volts) 
(Eq. 16) 
Where the square root converts the IIM2 powers to voltages. The terms iim2, iim2
_{1}, and iim2
_{2} are in linear power units (watts or milliwatts). Also, G
_{1} (dB) = 10 × log
_{10}(g1) and IL (dB) = 10 × log
_{10}(il).
Equation 15 can be rearranged into the following equation:
IIP2 = P_{I} + (P_{I}  IIM2) (dBm) 
(Eq. 17) 
Equation 17 defines the input IP2 for the whole receiver, and can be rewritten in linear power units (milliwatts, or mW) instead of dBm:
p_{I}/iip2 = iim2/p_{I} 
(Eq. 18) 
Similar to what we did with equation 17, we can define IIP2
_{1} and IIP2
_{2} for block 1 and block 2, respectively, as:
IIP2_{1} = P_{1} + (P_{1}  IIM2_{1}) (dBm) 
(Eq. 19) 
IIP2_{2} = P_{2} + (P_{2}  IIM2_{2}) (dBm) 
(Eq. 20) 
Knowing that P
_{1}(dBm) = P
_{I} and P
_{2}(dBm) = P
_{I} + (G
_{1}  IL  S), we obtain from equations 19 and 20:
IIP2_{1} = P_{I} + (P_{I}  IIM2_{1}) (dBm) 
(Eq. 21) 
IIP2_{2} = (P_{I} + G_{1}  IL  S) + (P_{I} + G_{1}  IL  S  IIM2_{2}) (dBm) 
(Eq. 22) 
As we have done for equation 17, equations 21 and 22 can be rewritten in linear power units instead of dBm to result in equations 23 and 24, respectively:
p_{I}/iip2_{I} = iim2_{I}/p_{I} 
(Eq. 23) 
(p_{I} × (g_{1}/il))/(iip2_{2} × S²) = iim2_{2}/(g_{1}/il) × p_{I}) 
(Eq. 24) 
Where S(dB) = 10 × log
_{10}(s) and IL(dB) = 10 × log
_{10}(il). Note that S(dB) and IL(dB) are positive numbers.
Returning to equation 16 and dividing it by (p
_{I})
^{1/2},we get equation 25:
√(iim2/p_{I}) = √(iim2_{I}/p_{I}) + √(iim2_{2}/(g_{1}/il) × p_{I}) 
(Eq. 25) 
If we substitute each term in equation 25 by equivalent terms using equations 18, 23, and 24, and after simplifying by removing the term p
_{I}, we obtain the following modified IIP2 cascading equation:
√(1/iip2) = √(1/iip2_{I}) + √((g_{1}/il)/(iip2_{2} × s²) 
(Eq. 26) 
As we can see from equation 12, by choosing a highly selective RF filter (s >> 1), we can minimize the effect of the first mixerblock input IP2 (IIP2
_{2}) on the overall receiver RF frontend input IP2 (IIP2). It is worth noting that in a cascaded system analysis the first mixerblock input IP2 (IIP2
_{2}) should be replaced with an equivalent input IP2. This exchange effectively introduces selectivity in the RF filter. This equivalent IIP2
_{2} can be written as:
IIP2^{e}_{2} = IIP2_{2} + 2 × S (dBm) 
(Eq. 27) 
A more generalized equation to predict the overall input IP2 of a receiver's RF frontend chain with M cascaded stages can be formulized based on equation 26. Each stage has a linear gain (g
_{n}), an input IP2 (iip2
_{n}, in watts), and a selectivity (s
_{n}) at the halfIF CW tone frequency. This introduces an inband IM2 product (assuming that il
_{n} << s
_{n}):
√(1/iip2) = √(1/iip2_{1}) + √(g_{1}/(iip2_{2} × s_{1}²)) + √((g_{1} × g_{2})/(iip2_{3} × (s_{1} × s_{2})²) + ... + √((g_{1} × g_{2} ... g_{M1})/(iip2_{M} × (s_{1} × s_{2} ... s_{M1})²) 
(Eq. 28) 
Where S
_{n}(dB) = 10 × log
_{10}(s
_{n}).
References
 S. Maas, Microwave Mixers, Norwood, MA, Artech House, 1993.
 P. Vizmuller, RF Design Guide, Norwood, MA, Artech House, 1995.
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APP 749: May 01, 2002
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Appnote749,
Appnote 749
